1. Field of the Invention
The present invention relates to power converters, and more particularly to half-bridge resonant converters.
2. Description of the Related Art
In recent years, miniaturization and power saving have become important issues in electronic equipments, especially in portable electronic equipments, and hence power converters used in electronic equipments are required to reduce the size and weight and increase the power conversion efficiency. This is achieved by using switching-mode power converters with semiconductor power switches. In reducing power converter size, designers have turned to increased switching frequencies. Higher switching frequencies allow for smaller, lighter inductive/capacitive energy storage and filter components, but also bring with them increased switching losses.
Switching losses include the power loss which results from the simultaneous presence of voltage and current in the semiconductor switch during turn-on and turn-off transitions. The semiconductor switch, for example, may be implemented by metal-oxide-semiconductor field-effect transistor (MOSFET). In addition, switching losses further include the power loss which results from the charging and discharging of the parasitic capacitance across the drain and source of the MOSFET switch. As the switching frequency increases, so do the switching losses. Excessive switching losses can result in damage to the switch and poor power conversion efficiency.
In order to reduce switching losses, resonant concepts are applied to switching power converters to allow zero-voltage switching (ZVS) and/or zero-current switching (ZCS) so as to minimize switching losses. All resonant converters operate in essentially the same way: a square pulse of voltage or current is generated by the power switches and this is applied to a resonant circuit. Energy circulates in the resonant circuit and some or all of it is then tapped off to supply the output.
Referring to FIG. 1, it is a schematic diagram of a conventional half-bridge series resonant converter 100. Power provided by a DC power source 200 is delivered to the resonant converter 100 at an input voltage VIN and is delivered to a load 300 at an output voltage VOUT. The resonant converter 100 includes a half-bridge switching circuit 110 for alternatively coupling two terminals of the DC power source 200 to an input of a resonant circuit 120. The resonant circuit 120 includes resonant inductor LR and resonant capacitor CR coupled in series. In general, the resonant inductance LR may comprise, in whole or in part, the leakage inductance of the transformer 130. The rectifier circuit 140 includes diodes D3, D4 and an output capacitor CO to generate the output voltage VOUT from an output of the resonant circuit 120.
In FIG. 1, the switching circuit 110 includes a half-bridge circuit 112 with two switches Q1, Q2 controlled by a control circuit 114. The switches Q1, Q2 are implemented by MOSFET. As switching current reversal is required, all switches must have freewheeling diodes. In the switching circuit 110, the switches Q1, Q2 employ external freewheeling diodes D1, D2. The control circuit 114 provides 50% duty cycle symmetrical control signals, which are identical to each other with 180-degree phase shift, to drive the switches Q1, Q2 in a substantially complementary fashion such that the two terminals of the DC power source 200 are alternatively coupled to the input of the resonant circuit 120. In practice, a short dead time should be set up to avoid the simultaneous conduction (or turn-on) of the two switches; hence, the duty cycle of the two control signals is not 50% but generally close to 50%. During the dead time, both of the switches Q1, Q2 are turned off.
Referring to FIG. 2, it is a timing diagram of the control signals of the switches Q1, Q2 and voltages across the switches Q1, Q2 in the resonant converter 100 of FIG. 1 for full load or light load. In the present embodiment, the control signals are two gate-to-source voltages VGS1, VGS2, and the voltages across the switches Q1, Q2 are two drain-to-source voltages VDS1, VDS2. Here, the dead time τD is exaggerated for display. It is obvious that the switch Q1 is turned on when the voltage VDS1 across the switch Q1 is zero, and the switch Q2 is turned on when the voltage VDS2 across the switch Q2 is zero. In other words, both switches Q1, Q2 are operated under ZVS so as to minimize switching losses. In such resonant converter 100, as the load 300 is changed, the output voltage VOUT (or output power) muse be regulated by controlling the switching frequency fS (or the reciprocal of the switching period T).
Referring to FIG. 3, it is a diagram of output power v.s. frequency of the resonant converter 100 of FIG. 1. Under a light load (see curve 31), the resonant converter 100 operates at the operating point 31′ whose frequency is above and further away from the resonant frequency fR1. When the load becomes heavy (see curve 32), the resonant converter 100 must provide more output power to drive the load, hence it operates at the operating point 32′ whose frequency is above and closer to the resonant frequency fR2. The dashed curve 33 is the locus of operating points. It is shown that the switching frequency (or frequency of operating point) is changed as load is changed. The characteristic of variable (switching) frequency operating has the disadvantage of making the control and the electromagnetic interference (EMI) filter design more complicated.